A TDR (time-domain reflectometer) which requires no oscilloscope and no expensive parts. A TDR built out of mostly cheap components. Bandwidth about 2.5 GHz, time step 20 ps (can be set down to about 1.5 ps). Has averaging, autocalibration, firmware which guides the user through the calibration and measurement and software for PC which allows more comfortable user interface and calibration.
Also, it is my diploma thesis.
The reflectometer is able to measure in 20 ps steps (or any other step, it is just a matter of software configuration). The pulse generator makes rectangular wave with risetime of 85 ps. Measured risetime is about 200 ps. As a standalone device, it can detect simple discontinuities, such as shorts, disconnected cable, a split which halves impedance. When connected to PC, it can perform OSL calibration and show calibrated data. If you want to know more, look into project logs.
Last time, we've got covered the method for sampling using diodes. However, that's only the beginning of sampling. Then you need to sample into something and then use it for something. Of course, it has to be a capacitor or something capacitor-like. This means you can sample into capacitor, gate of field-effect transistor, input of unipolar op-amp etc. I already mentioned the Houten's sampling head, which uses input of TL072 as sampling capacitor. However, its capacity is about 15 pF. This would heavily load the sampling bridge and also the input of the sampling head. Therefore the sampling head uses another 1 pF capacitor in series with the op-amp, which forms a capacitive divider and also limits loading the sampling bridge.
There is a reason why use a unipolar amplifier. You have to store the sampled voltage, but that's only one part of the job. After storing the voltage, you want to read it, probably. Well, unless you are Schrödinger's relative and don't actually want to know the voltage. Speaking of Schrödinger, that is the problem. Reading the voltage will probably somehow alter the sampled voltage. If you use bipolar amplifier after the sampling circuit, you will experience an effect calleddroop. Because a bipolar amplifier has a significant input current (remember, the sampling capacitor is only a few picoFarads), the sampling capacitor gets gradually charged (or discharged). If the time between sampling the voltage and measurement is always the same, you will obtain the data DC shifted, downscaled or mix of both. If the time between those events is not constant, the data will look as if they were noisy. The noise is approximately equal to the droop rate multiplied by the jitter of the time between sampling and measurement (if the droop is approximately linear).
Therefore there are two things you want to avoid. Input current of amplifier after the sampling circuit should ideally be zero and the time between sampling and measurement of the sampled voltage should be constant. And as short as possible.
This leads us back to the unipolar amplifiers. These have almost no input current, or at least negligible if the time between sampling and measurement is short enough. You can use MOSFETs, most JFETs, MOSFETs and most unipolar op-amps. However, avoid HEMTs, even if they are called field effect, since their input current is almost the same as in bipolar transistors. Or at least this applies to those which have a proper datasheet (many have only a few limiting values and a set of S-parameters).
To fight with droop, my reflectometer uses BF998 MOS tetrode as input amplifier followed by second sampler section with much larger sampling capacitor (1 nF). These two samplers are precisely synchronised. The output of the second sampler is then measured by ADC. There is one slight error, the C231 in source of the tetrode is not used in the final reflectometer. It would make the amplifier unstable or make it oscillate. The same would happen if the C230 was larger, R251 was smaller and so on. The circuit is tuned to be as fast as possible while being stable. Making the C230 larger also loads the sampling bridge much more and causes the input impedance of the instrument drop at higher frequencies. Since the bonding wires of the sampling bridge are inductive, the sampling bridge forms a resonator with the C230.
To explain why there are so many resistors and capacitors placed in seemingly strange places, have a look at the transfer function of the amplifier. Please note that the -6 dB corner frequency is predicted by simulation to be at about 4 GHz, while the transistor is recommended for applications under 1 GHz. Yes, it smells of some positive feedback. But carefully crafted not to turn into instability. Sorry for the captions in graph being in Czech, the bright trace is S21 parameter.
The transfer function of the whole sampler, from test connector to the output of the BF998 should look somewhat like the next image. According to...
Have you ever had the need to sample some analog signal? If so, it was probably at low frequencies. Quite probably some audio grade stuff. Everyone knows how to do that. Either take a complete sample-and-hold amplifier like the old LF398 or the DIY way by connecting together some generic amplifier, capacitor and some electronically controlled switch like the 4066.
But what if you look a bit higher? Several megahertz and up? Okay, the 4066 method still may be useable. You can find quite a lot of receivers which use a 4053 or 4066 as an IQ demodulator for short-wave amateur radio. Still good, still cheap and simple.
But the true question is what happens once you want to sample something which extends beyond several hundred megahertz or even several gigahertz. Then you will find yourself in a very uncomfortable place. If you wish to do some voodoo with OTAs, you can reach hundreds of megahertz, but for more you have to dive into your pocket and look very deep in your pocket. First, you will be shocked because there are only a few parts which can do such things, like the HMC760. But seriously, are you going to pay $360 for a chip? Maybe, but not me.
Okay, so searching at e-shops answered the question showed us there is no part which would do this job for us. Time for diving into old manuals, schematics and articles! Maybe some of you have already met some old TDR machine or a 60s Tektronix sampling oscilloscope. They usually have quite similar sampler. Some are very simple, some are very intricate. But the principle is always the same - using diodes as sampling bridges. When current passes through them, so does signal. Turn off the current and the signal also stops going through. Ideally, that is.
What you can see on this part of schematic form S-2 sampling head is the sampler. It doesn't look obvious, does it? If you don't believe it is a sampler, read the manual which explains the whole sampling head and each of its parts, how it works and how to make it work. It's sad you don't get a manual like this these days. Note that this sampling head was introduced in 1967 and offered 4.6 GHz of bandwidth and <75 ps of risetime.
To summarize it: a very short pulse of current of defined length goes through the diodes, turns them on briefly and then turns them off again very fast. The two key parameters here are how fast you can turn off the diodes and how precisely you can control at what time it happens. Let's look at how the folks at Tek did this magic.
What you see is an avalanche pulse generator. You may have already seen it in one of my projects. Essentially it is transistor which has collector-emitter voltage so high it almost goes into avalanche breakdown, but not on its own. The avalanche is triggered by other circuitry using the transformer on the right. This makes a very fast pulse. To make it even faster, a snapoff diode is used (these are very hard to come by these days). The length of the pulse is precisely set by the two "cliplines" on the left.
It is fast and great, but it requires several supply voltages (and quite high!), draws a lot of current and cannot be triggered very often, because it has to reach steady state again. What about doing it less complicated?
If you look at article by Hubert Houtman, you can see that it is possible to achieve about 1 GHz of bandwidth using just a "quite" fast comparator and two resistors.
The MAX961 comparators have rise/fall times about 2.3 ns. The outputs of the Si5351 have those times under 1 ns and are implemented as current sources. Theoretically, this could get us at least 2 GHz of bandwidth using the same diodes. Since we can get faster diodes today, it could be even better. The only unpleasant side effect is a need for a virtual analog ground at 1.65 V potential (half the 3.3 V supply for the Si5351).
To see how simple it can be, look at the lower right part of the next image. Four diodes, three resistors and directly connect it to Si5351. Nothing complicated. The result is about 200...
When I was designing this TDR, I was trying to avoid using expensive parts. Thus, ECL delay lines were out of question. I found a few articles which were using other ways to achieve equivalent-time sampling.
For example, there was a project which used two oscillators which were slightly out of tune. The problem was that it was highly "academic". The time step was dependent on the frequency of those two free-running oscillators, thus it depended on temperature, supply voltage and moon phase. Good idea, but not useable in such form. See the original article on IEEE XPlore (or look it up using SciHub or Library Genesis if you do not have access to scientific articles) A 16ps-resolution Random Equivalent Sampling circuit for TDR utilizing a Vernier time delay generation.
There were also several projects which used FPGA which implemented a DDS. This DDS generated a sinewave whose phase could be arbitrarily set in very small steps. Then this sinewave was turned into square wave. This way, the researchers were able to generate pulses for generating a pulse for the TDR and sampling pulses in one FPGA and a handful of components. The sampling was performed by a fast latchable comparator. In my opinion, this is the best architecture of TDR which I have seen in scientific articles, however I wanted to avoid FPGAs. See Miniaturized FPGA-Based High-Resolution Time-Domain Reflectometer
And then there was the usual bag of sh** which you can find in scientific articles, usually from universities which have to publish articles, even though they have nothing to show. This usually revolves around connecting pulse generator to oscilloscope, several pages of description of the splitter used for connecting together the generator, oscilloscope and DUT, then several pages covered in equations describing everything from Maxwell equations to the effect of solar eclipse on price of donuts after Easter 1919.
And so it seemed I had to come up with something new (almost). I was playing with dual-VCO PLLs Si5351 at the time, so I thought if I could use them for this purpose. The answer was "probably yes, why not try it". And so I did. These PLLs allow you to set frequency of the two VCOs which can be non-integer. The fractional part allows the two frequencies to differ by less than 1 ppm. And the relation is precisely controlled as opposed to the "Vernier" article. So, nothing really new, just using newer and more interesting parts. When set to exactly 1 ppm difference in frequency, one can get one million of samples during one measurement cycle. I set the measurement cycle to 10 microseconds so it could be easily measured by internal ADC of STM32 microcontroller. This gives the ability to measure 1 million of samples 10 picosecond apart.
The frequency of the two VCOs is below, parameters a, b and c are the integer and fractional part of multiplier, d is the divider.
The time step of sampling is then given by the next equation:
From these equations, you can find out how to set the parameters to suit your needs (time step, number of points, measurement length).